Fiber optic digital transmission system

ABSTRACT

A transmission system responsive to an electrical input information signalncludes means for modulating the input signal with a microwave subcarrier signal to produce a subcarrier modulated signal. A first interferometric modulator responsive to a source light and to the subcarrier modulated signal produces a first modulated optical signal. A second interferometric modulator cascade coupled to the first modulator and being responsive to the first modulated optical signal and to a microwave local oscillator signal produces a second modulated optical signal. Means coupled between the first and second modulators transmit the first modulated optical signal from the first modulator to the second modulator. Means responsive to the second modulated optical signal produce an output signal.

CROSS REFERENCE TO RELATED PATENTS AND APPLICATIONS

This application is a Continuation in Part of commonly assigned and U.S.application Ser. No. 08/258,028, now U.S. Pat. No. 5,422,966 filed Jun.10, 1994 by the inventors Ganesh K. Gopalakrishnan and William K. Burnsand having docket no. NC 75,680 and is entitled to the benefit of theJun. 10, 1994 filing date for the matter disclosed therein. U.S.application Ser. No. 08/258,028, now U.S. Pat. No. 5,422,966 isincorporated herein by reference.

BACKGROUND OF THE INVENTION

The present invention relates to fiber optic links for signaltransmission, and more particularly, to microwave subcarrier multiplexedsystems.

DESCRIPTION OF THE RELATED ART

A mixer is a nonlinear device which combines signals of differentfrequencies to produce a common output signal having the sum ordifference. Typically, a local oscillator (LO) reference signal isapplied to one of the mixer ports, and a weaker signal (RF) to beconverted is applied to a second mixer port. Sum or difference (IF)frequencies of the input signals and, usually, other higher orderharmonics appear at the output port of the mixer. For transmitterapplication, an information bearing signal is upconverted through amixer to a higher RF microwave frequency and is usually amplified beforebeing transmitted. This up-conversion usually involves the sumfrequency. For receiver application, a received microwave signal isdown-converted through a mixer to a lower IF frequency, which is thendemodulated.

In conventional optoelectronic receivers, RF signals transmitted over anoptical fiber are detected with a photoreceiver, following whichdown-conversion to an IF signal is accomplished through a conventionalmicrowave mixer. However, there are several difficulties associated withthis approach. If the transmitted optical signal is modulated atmillimeter wave frequencies (the carrier frequency), then a high-speedphotoreceiver is necessary to detect the transmitted signal. Thephotoreceiver must have high enough bandwidth to detect the carriersignal. High-speed receivers, because of their fine geometries,typically saturate at fairly low optical power levels. These low powerlevels may be insufficient to overcome the inherent noise level of theoptoelectronic system. Conventional microwave mixers have a limitedbandwidth and are limited in port-to-port isolation. Such mixers alsohave high intermodulation (IM) distortion. Some of these problems can becircumvented by optical down-conversion, thereby eliminating the needfor high-speed detection and the microwave mixer.

U.S. Pat. No. 5,199,086 to Johnson proposes an electro-optic mixersystem using two electro-optic interferometric waveguide modulatorscoupled in series. However, the electro-optic modulators in the Johnsonsystem may have high drive voltage. As a result, the mixer systemproposed in Johnson would have high insertion, i.e. conversion loss andthus low efficiency. Such a system is not practical because, as with theconventional optoelectronic receivers, the modulation effect in themodulated optical signal would be masked by noise.

Transmission of digital signals over an optical fiber offers theadvantage of low-loss and lack of electromagnetic interference, andseveral approaches for fiber optic links have been proposed. However,one such approach, resonant enhancement of semiconductor lasers,provides a system of narrow bandwidth, and another such approach,phase-locked distributed-feedback laser systems, has limited bittransmission rates.

SUMMARY OF THE INVENTION

It is an object of this invention to provide a fiber optic link for datatransmission using microwave subcarrier multiplexing.

It is a further object of this invention to provide a fiber optic linkfor digital data transmission using microwave subcarrier multiplexing.

It is another object of the invention to provide a fiber optic link fordigital data transmission using subcarrier multiplexing that canconcurrently detect and down-convert microwave signals using low-speedphotoreceivers.

Another object of the invention is to provide a fiber optic link fordigital data transmission using subcarrier multiplexing and externalmodulation.

These and other objectives are achieved by a transmission systemresponsive to an electrical input information signal which includesmeans for modulating the input signal with a microwave subcarrier signalto produce a subcarrier modulated signal. A first interferometricmodulator responsive to a source light and to the subcarrier modulatedsignal produces a first modulated optical signal. A secondinterferometric modulator cascade coupled to the first modulator andbeing responsive to the first modulated optical signal and to amicrowave local oscillator signal produces a second modulated opticalsignal. Means coupled between the first and second modulators transmitthe first modulated optical signal from the first modulator to thesecond modulator. Means responsive to the second modulated opticalsignal produce an output signal.

These and other objects, features and advantages of the presentinvention are described in or apparent from the following detaileddescription of preferred embodiments.

BRIEF DESCRIPTION OF THE DRAWINGS

The preferred embodiments will be described with reference to thedrawings, in which like elements have been denoted throughout by likereference numerals, and wherein:

FIG. 1 shows in block diagram form an electro-optic mixer of the presentinvention.

FIG. 2A illustrates a coplanar strip (CPS) electrode structure.

FIG. 2B illustrates a coplanar waveguide (CPW) electrode structure.

FIG. 3 illustrates dispersion curves for substrate modes of Z-cutlithium niobate (LiNbO₃) slabs or substrates.

FIG. 3A illustrates an exemplary Z-cut LiNbO₃ substrate from which wasderived the waveform for the TM₀ (cond) substrate mode shown in FIG. 3.

FIG. 3B illustrates an exemplary X- or Y-cut LiNbO₃ substrate.

FIGS. 4A and 4B show electrical transmission (S₂₁) plotted againstfrequency for different substrate thicknesses of CPS devices.

FIGS. 4C and 4D show electrical transmission (S₂₁) plotted againstfrequency for different substrate thicknesses of CPW devices.

FIG. 5 illustrates a cross section of a phase modulator with coplanarwaveguide electrodes.

FIG. 6 illustrates a top view of the phase modulator of FIG. 5.

FIG. 7 illustrates a cross section of an intensity modulator withcoplanar waveguide electrodes on a Mach-Zehnder interferometer.

FIG. 8 illustrates a top view of the intensity modulator of FIG. 7.

FIG. 9 illustrates a plot of the microwave index n_(m) against electrodethickness, and a comparison of theoretical results with experimentalresults.

FIG. 10A illustrates the electrical transmission through coplanarmicrowave waveguides.

FIG. 10B illustrates the normalized optical response of the device ofFIG. 5 or FIG. 7 in decibels (dB) of electrical power as a function offrequency.

FIG. 11 shows in block diagram form an embodiment of the electro-opticmixer of FIG. 1 used in the examples herein.

FIG. 12 illustrates the optical response and the RF half-wave voltagefor modulators M₁ and M₂ of FIGS. 1 and 11.

FIG. 13 shows the power spectrum of the mixer of FIG. 11 foroff-quadrature bias.

FIG. 14 shows the power spectrum of the mixer of FIG. 11 for bias atquadrature.

FIG. 15 shows the variation of detected IF power with respect to inputRF power for the mixer of FIG. 11 when RF=15.52 GigaHertz (GHz) andLO=15.56 GHz.

FIG. 16 shows the variation of detected IF power with respect to inputRF power for the mixer of FIG. 11 when RF=40.02 GHz and LO=40.06 GHz.

FIG. 17 shows the variation of detected IF power with respect to inputRF power for the mixer of FIG. 11 for equal strength RF and LO signals.

FIG. 18 illustrates the relation between RF Half-Wave Voltage andconversion loss.

FIG. 19 illustrates a mixer employing an erbium doped fiber amplifierbetween the first and second modulators.

FIG. 20 illustrates a mixer employing an erbium doped fiber amplifier toamplify the output optical IF signal.

FIG. 21 shows a fiber optic transmission system of the presentinvention.

FIG. 22 shows the error rate of the device of FIG. 21.

FIG. 23 shows a fiber optic transmission system using multiplesubchannels.

FIG. 24 shows a fiber optic transmission system including three opticalmodulators.

FIG. 25 shows a fiber optic transmission system using multiplesubchannels.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

Referring now to the drawings, FIGS. 1 and 11 show an electro-opticmixer 10. The mixer 10 includes means 20 for producing first and secondmicrowave signals S₁ and S₂ having first and second frequencies f₁ andf₂, respectively. In other words, the first and second microwave signalsS₁ and S₂ have first and second angular frequencies ω₁ and ω₂,respectively. A device 30 provides source light L_(i). The device 30preferably includes a cw laser 32, polarizing optics 34, and apolarization-preserving optical fiber 36 for producing, preferably,polarized light L_(i), and most preferably, polarized cw laser lightL_(i).

The mixer 10 also includes first and second broadband, low drivevoltage, traveling wave intensity modulators M₁ and M₂, respectively,which are described further below. The first modulator M₁ is responsiveto the source light L_(i) and to the first microwave signal S₁ forproducing a first modulated optical signal L_(m1). The mixer 10 furtherincludes means 40 coupled between the first and second modulators M₁ andM₂, respectively for transmitting the first modulated signal L_(m1) fromthe first modulator M₁ to the second modulator M₂. Since the modulatorsM₁ and M₂ are electrically decoupled, the isolation between their signalports is almost infinite, making them attractive for microwave mixing.The transmitting means 40 is preferably, but not necessarily,polarization-maintaining means 40, such as polarization-maintainingoptical fiber 40. The second modulator M₂ is cascade coupled to thefirst modulator M₁ and responsive to the first modulated signal L_(m1)and to the second microwave signal S₂ for producing a second modulatedoptical signal L_(m2) having a frequency component at the difference |f₁-f₂ | between the first and second frequencies f₁ and f₂, respectively.Such an embodiment of the mixer 10 would be useful for down-conversion.Alternatively, the second modulator M₂ is cascade coupled to the firstmodulator M₂ and responsive to the first modulated signal L_(m1) and tothe second microwave signal S₂ for producing a second modulated opticalsignal L_(m2) having a frequency component at the sum f₁ +f₂ of thefirst and second frequencies f₁ and f₂, respectively. Such an embodimentof the mixer 10 would be useful for up-conversion.

EXEMPLARY MODULATORS M₁ AND M₂

Referring now to FIGS. 2A and 2B, FIG. 2A illustrates an exemplarycoplanar strip (CPS) electrode structure in an exemplary Mach Zehnderinterferometer modulator, while FIG. 2B illustrates an exemplarycoplanar waveguide (CPW) electrode structure in an exemplary MachZehnder interferometer modulator.

Physically the CPS electrode structure of FIG. 2A has a hot electrode112, to which a modulating signal is applied (not shown), and a singleground plane or grounded electrode 114 on one side of the hot electrode112. On the other hand, the CPW electrode structure of FIG. 2B has a hotcentral electrode 112A, to which a modulating signal, typically amicrowave signal such as S₁ or S₂ (not shown) is applied, and two groundplanes or grounded electrodes 114A and 114B on opposite sides of the hotcentral electrode 112A. The major physical difference between the CPSand CPW electrode structures is that the CPS electrode structure of FIG.2A lacks a second ground plane or grounded electrode. This physicaldifference results in a crucial operational difference between the useof the coplanar strip (CPS) electrode structure of FIG. 2A and the useof the coplanar waveguide (CPW) electrode structure of FIG. 2B in anintegrated optic modulator. The advantage of the CPW structure of FIG.2B over the CPS structure of FIG. 2A will be explained by now referringto FIG. 3 and FIG. 4A through 4D.

FIG. 3 illustrates dispersion curves for substrate modes of Z-cutlithium niobate (LiNbO₃) slab (shown in inserts 118A and 118B), whichLiNbO₃ slab represents a dielectric substrate 116, and where air (havingan index=1) is the material under the LiNbO₃ substrate 116. Twodifferent conditions are shown in FIG. 3. One condition is with a metalcoating 120 on the upper surface of the dielectric substrate 116, whilethe other condition is without the metal coating 120 on the uppersurface of the dielectric substrate 116. The effective index n_(m) ofthe guided modes of the Z-cut LiNbO₃ slab is plotted against the ratioof h/λ₀, where h is the substrate 116 thickness and λ₀ is the free spacewavelength.

Shown in FIG. 3 are five substrate modes, respectively labeled TE₀(die), TM₀ (cond), TE₁ (cond), TM₀ (die) and TE₁ (die), where:

TE₀ (transverse electric) means that the electric field is parallel tothe plane of the slab (and the magnetic field is perpendicular to theplane of the slab),

TM_(o) (cond) represents the TM₀ (cond) substrate mode for a Z-cutLiNbO₃ substrate 116,

TE₁ (cond) represents the TE₁ (cond) substrate mode for a Z-cut LiNbO₃substrate 116.

the subscripts 0 and 1 refer to the order of the substrate mode, with 0and 1 respectively representing the first order mode and the secondorder mode,

(die) represents an approximation of the CPS structure based on theassumption that substantially no metal is disposed on the top of thedielectric substrate 116,

(cond) represents the CPW structure where metal 120 (a conductor) isdisposed on the top of the dielectric substrate 116,

2.15=the optical index in LiNbO₃ at a wavelength of 1.3 micrometer (μm),

h=the thickness of the substrate 116,

λ₁ =free space microwave wavelength,

n_(m) =n_(m) (subs)=the effective index of substrate mode, and

n_(m) (CPW)=the effective index of the CPW coplanar mode (which forZ-cut LiNbO₃ =2.15 at 1.3 μm when optical-microwave phase match isachieved).

It has been known in the prior art that microwave leakage (loss dips intransmission) is due to a microwave coupling between the coplanar modeand a substrate mode bounded by the top and bottom surfaces of thesubstrate 116. In microwave terminology this is called a surface wave.However, for purposes of this description, the optical terminology of asubstrate mode will be used. Although this microwave coupling has beenknown in the microwave area, it apparently has not been appreciated inthe optical modulator area. The frequency at which the microwavecoupling, and thus the microwave loss, begins depends on the thicknessof the substrate 116 and the dispersion behavior of the substrate mode.It is here where there is a crucial difference between the use of thecoplanar strip (CPS) electrode structure of FIG. 2A and the use of acoplanar waveguide (CPW) electrode structure of FIG. 2B. It should berecalled that the CPW structure of FIG. 2B has ground planes 114A and114B on both sides of the central electrode 112A, whereas the CPSstructure of FIG. 2A has a ground plane 114 on only one side of theelectrode 112. The dispersion curves for the Z-cut LiNbO₃ substrate 116(shown in inserts 118A and 118B of FIG. 3) are different, depending onwhether the upper surface of a substrate 116 is coated with the metal 30(insert 118B) or is not coated with the metal 120 (insert 118A).

FIG. 3 shows that the TE₀ (die) substrate mode of the non-metal coatedsubstrate 116 of insert 118A occurs at a lower frequency than for theTM₀ (cond) substrate mode of the metal coated substrate 116 of insert118B. From FIG. 3, it can be determined that the coplanar mode-substratemode coupling problem can be solved by using a thick electrode CPWstructure (wherein the metal of the thick electrode CPW structuresubstantially coats the entire top surface) and by making the substratethickness sufficiently thin that microwave leakage will not occur withinthe bandwidth of interest. The required thickness of the electrodestructure for optical-microwave phase match can be calculated separatelyby known techniques (to be discussed later).

As stated before, the dispersion curves shown in FIG. 3 apply to thespecific cases where Z-cut slabs or substrates of LiNbO₃ are utilizedand air is the material underneath each of the LiNbO₃ substrates 116. Inthese cases the effective index (n_(m)) of the CPW coplanar mode isadjusted to be close to 2.15.

It will be recalled that microwave leakage (loss dips in transmission)is due to a microwave coupling between the coplanar mode and a substratemode bounded by the top and bottom surfaces of the substrate 116. Toavoid such microwave leakage it is important that the substratethickness h be sufficiently small at a given λ₀ so that the followingequation is satisfied:

    n.sub.m (subs)≦n.sub.m (CPW)                        (1)

which for Z-cut LiNbO₃ the n_(m) (CPW)≅2.15 at a wavelength of 1.3 μmwhen optical-microwave phase match is achieved.

Since h and λ₀ are known, h/λ₀ can be readily calculated and then theeffective index of the substrate mode n_(m) (subs) can be determinedfrom the dispersion curves in FIG. 3. For example, if h/λ₀ =0.04, thenthe effective index n_(m) (subs) of the TM₀ (cond) substrate mode forthe Z-cut LiNbO₃ substrate 116 would be about 2.5. Since n_(m)(CPW)=2.15 and n_(m) (subs)=2.5, n_(m) (subs) would not be ≦n_(m) (CPW)and therefore microwave losses would result when h/λ₀ =0.04. To preventsuch a microwave loss, h (the substrate thickness) would have to bethinned until the relationship shown in Equation (1) is satisfied acrossa desired bandwidth. When this relationship is satisfied, substantiallyno microwave leakage occurs across that desired bandwidth.

There are several steps that have to be taken to determine the geometryof the broadband, travelling wave, electro-optic integrated opticalmodulator of the present invention which contains the thick electrodecoplanar (CPW) structure on a thin substrate. After the optical index isdetermined:

1. The coplanar microwave index has to be made equal to the opticalindex (to be explained in discussion of FIG. 9), and

2. Then the substrate must be thinned so that the effective index (ormicrowave index) of the substrate mode is less than or equal to theeffective index (or microwave index) of the CPW coplanar mode. (SeeEquation (1).)

Referring to FIGS. 3A and 3B, it will now be explained how the properthickness can be determined for any suitable Z-cut substrate 119A andfor any suitable X- or Y-cut substrate 119B, with each of the substrates119A and 119B having a metal coating 120 on the upper surface of thesubstrate. For a more universal application, the following Equations (2)and (3) will also be used in this explanation instead of the dispersioncurves of FIG. 3.

Equations for Leaky Mode Loss

For the Z-cut case: (TM₀ (cond)): ##EQU1##

For the X- or Y-cut case: (TM₀ (cond)): ##EQU2## where: h=substratethickness

λ₀ =free space microwave wavelength

n₀ =ordinary index of substrate

n_(e) =extraordinary index of substrate

n_(m) =effective index of substrate mode

n₃ =index of material below substrate

Each of the substrates 119A and 119B respectively utilized in FIGS. 3Aand 3B is a uniaxial crystal which has two indices of refraction, onefor fields parallel to Z and the other for fields parallel to X or Y.The Equations (2) and (3) are for the two different orientations of theuniaxial crystal substrates shown in FIGS. 3A and 3B. More particularly,Equation (2) is for the Z-cut orientation shown in FIG. 3A, whileEquation (3) is for the X- or Y-cut orientation shown in FIG. 3B. Inaddition, Equation (2) determines the ratio h/λ₀ as a function of theeffective index n_(m) for the TM₀ (cond) substrate mode for the Z-cutsubstrate being used. In a similar manner, Equation (3) determines theratio h/λ₀ as a function of the effective index n_(m) for the TM₀ (cond)substrate mode for the X- or Y-cut substrate being used.

Assuming that Equation (2) is being utilized during the implementationof the Z-cut substrate for an electro-optic integrated optical modulatorof the invention, then the effective index of the CPW coplanar moden_(m) (CPW) would be known. In addition, the values h, λ₀, n₀, n_(e) andn₃ in Equation (2) would also be known. Only n_(m) or n_(m) (subs), theeffective index of the TM₀ (cond) substrate mode, would not be known. Byinserting the known values of h, λ₀, n₀, n_(e) and n₃ into Equation (2),the remaining value of n_(m) (subs) can be determined and compared withn_(m) (CPW), the effective index of the CPW coplanar mode, to see if therelationship shown in Equation (1) [n_(m) (subs)≦n_(m) (CPW)] issatisfied. If the relationship is satisfied, then the substrate is thinenough. If the relationship is not satisfied, then h must be thinneduntil Equation (1) is satisfied.

In a similar manner, if Equation (3) is being utilized during theimplementation of the X- or Y-cut substrate for an electro-opticintegrated optical modulator of the invention, then the effective indexof the CPW coplanar mode n_(m) (CPW) would be known. In addition, thevalues h, λ₀, n₀ and n₃ in Equation (3) would also be known. Only n_(m)or n_(m) (subs), the effective index of the TM₀ (cond) substrate mode,would not be known. By inserting the known values of h, λ₀, n₀ and n₃into Equation (3), the remaining value of n_(m) (subs) can be determinedand compared with n_(m) (CPW), the effective index of the CPW coplanarmode, to see if the relationship shown in Equation (1) [n_(m)(subs)≦n_(m) (CPW)] is satisfied. If the relationship is satisfied, thenthe substrate is thin enough. If the relationship is not satisfied, thenh must be thinned until Equation (1) is satisfied.

In an earlier experiment, a 10 μm thick electrode CPW structure and a 14μm thick electrode CPS structure were each fabricated on Z-cut LiNbO₃slabs or substrates 116 with a 0.9 μm thick SiO₂ buffer layer on thesubstrate. Hot electrode widths were 8 μm, gap widths between a hotelectrode and its associated ground plane(s) were 15 μm, and groundplanes were 2-3 millimeters (mm) wide. Finite element calculationsindicated that those geometries should result in microwave indices near2.4. These devices were fabricated on substrates 0.5 and 0.25 mm thickand 8 mm wide. The electrode interaction length was 2-4 cm.

The devices were tested on a Hewlett-Packard HP-8510C automatic networkanalyzer with the electrical transmission (S₂₁) shown in FIGS. 4A, 4B,4C and 4D. The CPS devices showed loss dips beginning at 7 GHz for thedevice with the 0.5 mm thick substrate and at 14 GHz for the device withthe 0.25 mm thick substrate. The CPW devices showed similar behavior atabout 24 GHz for the device with the 0.5 mm thick substrate, but nosignificant dips at all out to 40 GHz for the device with the 0.25 mmthick substrate. The frequency at which mode coupling begins is referredto as f_(c). For frequencies above f_(c), it was verified that microwavepower was transmitted to the outer edge of the substrate (parallel tothe axis of the device) by perturbing the field at the edge with anotherslab of LiNbO₃, or by sawing off the edge and thus reducing the width ofthe substrate. These actions had the effect of changing the position ofthe loss dips with frequency but did not significantly affect f_(c).Power was observed at the outer edge of the substrate all along thelength of the device. The experimental results (Experiment) for f_(c)for several devices of each type are listed in the following TABLE,along with computed (Model) values from FIG. 3 assuming n_(m) =2.4.

                  TABLE                                                           ______________________________________                                        SUMMARY OF COMPUTED AND MEASURED                                              FREQUENCIES                                                                                      f.sub.c (GHz)                                              Device                                                                              h (mm)    Substrate mode                                                                             Model Experiment                                 ______________________________________                                        CPW   0.5       TM.sub.0 (cond)                                                                            25    24-26                                            0.25      TM.sub.0 (cond)                                                                            49    >40                                        CPS   0.5       TE.sub.0 (die)                                                                             11    7-8                                              0.25      TE.sub.0 (die)                                                                             22    14-16                                      ______________________________________                                    

Thus, as shown in FIGS. 4A and 4C (or FIGS. 4B and 4D) and the TABLE,for the same thickness substrate the CPS structure shows in FIG. 4A(FIG. 4B) microwave leakage (loss dips in transmission) at lowerfrequencies than the CPW structure shows in FIG. 4C (FIG. 4D).

FIGS. 5 and 6 respectively show cross-sectional and top views of a highspeed phase modulator with coplanar electrodes.

In the phase modulator of FIGS. 5 and 6, a coplanar waveguide (CPW)structure 111, comprised of a center electrode 113 and ground planes orgrounded electrodes 115 and 117 on both sides of the center electrode113, is disposed on a thin substrate 119 of Z-cut lithium niobate(LiNbO₃) to avoid electrical leakage. Preferably the substrate 119 has athickness of from 0.16 to 0.24 mm and a width of about 8 mm.

The electrodes 113, 115 and 117 are preferably made of gold and havethicknesses of from 10-20 μm and electrode 113 has an exemplary width ofsubstantially 8 μm. The gap width G between the center electrode 113 andeach of the grounded electrodes 115 and 117 is selected to be about 15μm, while the grounded electrodes 115 and 117 are selected to be about2-3 mm wide.

The substrate 119 has electro-optic effects, and is coated with anexemplary silicon dioxide (SiO₂) buffer layer 121 having an exemplarythickness of substantially 0.9 μm. In addition, the substrate 119contains an optical waveguide 125 underneath the electrode 113. Theoptical waveguide 125 is formed by depositing a strip of titanium (Ti)metal on the surface of the LiNbO₃ substrate 119 and diffusing it intothe LiNbO₃ substrate 119 at high temperature by techniques well known inthe art. This is done before the SiO₂ buffer layer 121 and theelectrodes 113, 115 and 117 are deposited. Portions of the electrodes113, 115 and 117 extend in parallel paths over an electrode interactionregion of length L (to be explained) which is parallel to the opticalwaveguide 125. The silicon dioxide buffer layer 121 isolates the opticalfield in the optical waveguide 125 from the metal electrodes 113, 115and 117 of the coplanar waveguide structure 111 to prevent optical loss.

It should be noted that the phase modulator of FIGS. 5 and 6 wasfabricated with the above-specified geometries for operation at anexemplary 1.3 μm. Calculations have indicated that these above-specifiedgeometries should result in microwave indices n_(m) between about 2.2and about 2.4, as shown in FIG. 9 (to be explained).

In operation, 1.3 μm light from a light source 127, such as a laser, isfocused by a lens 129 into the optical waveguide 125 and propagatesthrough the optical waveguide 125. At the same time, a modulatingmicrowave drive signal, at an arbitrary amplitude of up to 4 to 5 voltspeak and at a frequency in the range from 0 Hz up to substantially 40GHz, is applied from a microwave source 131 to the coplanar waveguidestructure 111 (between an input port 152 on the center electrode 113 andeach of the grounded electrodes 115 and 117) and on the same side of theoptical modulator as the exemplary 1.3 μm light is transmitted into theoptical waveguide 125. The center electrode 113 also includes an outputport 153 for termination of the coplanar waveguide structure 111. Thelow drive voltage of up to 4 to 5 volts peak results in a highlyefficient optical modulator. This modulating drive signal modulates thephase of the propagating light or optical wave at the frequency of themicrowave drive signal. More particularly, the optical phase modulationresults from an interaction between the optical wave in the opticalwaveguide 125 and the microwave drive signal in the coplanar waveguidestructure 111.

The bandwidth of the phase-modulated optical signal is typically limitedby optical-microwave phase mismatch (wherein the microwave wave and theoptical wave travel at different velocities, depending on the design ofthe device). The phase matching or phase velocity matching in theinvention will now be discussed.

Since the microwave drive signal and the light waves are both applied onthe same side of the optical modulator, the microwave signal and thelight waves propagate in the same direction through the electrodeinteraction region of the modulator. The direction of propagation forboth the light waves and the microwave field is into the paper in FIG.5, and from left to right in FIG. 6.

The coplanar waveguide structure 111 is designed with theabove-discussed geometries so that the microwave field and the lightwaves propagate with substantially the same phase velocity when thelight has a wavelength of about 1.3 μm. When the microwave field and thelight waves have the same or matching phase velocities, the phasemodulator of FIGS. 5 and 6 has the best broadband responsecharacteristics.

The phase velocity match is determined by the effective indices of theoptical and microwave field modes. The effective index of the opticalmode is fixed by the index of refraction of the substrate 119. Theeffective index of the microwave field in the coplanar waveguidestructure is determined by the geometry of the electrode structure ofthe optical modulator of FIGS. 5 and 6, the width W of the centerelectrode 113, the gap G between the center electrode 113 and each ofthe grounded electrodes 115 and 117, the height H and geometry of theelectrodes 113, 115 and 117, the material and the dielectric constantsof the buffer layer 121 and the substrate 119, and the thickness of thesilicon dioxide buffer layer 121. These parameters are selected to makethe effective index of the microwave field mode as close as possible tothe effective index of the optical mode.

Of particular importance in producing a phase velocity match in theinvention is the use of a thick electrode structure (of from 10-20 μm)on a coplanar waveguide (CPW) structure 111 (wherein the metal of whichsubstantially coats the entire top surface of the buffer layer-coatedsubstrate 119, as indicated in FIGS. 5 and 6) and the use of a substrate119 having a thickness sufficiently thin (preferably of from 0.16 to0.24 mm) so that microwave leakage will substantially not occur withinthe bandwidth of interest (up to 40 GHz).

Referring now to FIGS. 7 and 8, FIGS. 7 and 8 respectively showcross-sectional and top views of a high speed intensity modulator withcoplanar electrodes. This design is the exemplary design for theintensity modulators M₁ and M₂ used in the mixer 10 of FIG. 1. FIGS. 7and 8 are respectively substantially the same as FIGS. 5 and 6, exceptthat they are cross-sectional and top views of a high speed Mach Zehnderinterferometer modulator instead of just the phase modulator of FIGS. 5and 6. The Mach Zehnder interferometer modulator of FIGS. 7 and 8produces an intensity modulation at its output.

The Mach-Zehnder interferometric modulator of FIGS. 7 and 8 isfabricated in a manner similar to the fabrication of the phase modulatorof FIGS. 5 and 6. More particularly, the Mach-Zehnder interferometer isused with a coplanar waveguide (CPW) structure 111 on a thin substrate119 of z-cut LiNbO₃ to avoid electrical leakage. The thickness of goldelectrodes 113, 115 and 117 are varied from 10-20 μm. The substrate 119is coated with a 0.9 μm SiO₂ buffer layer 121. The widths of theelectrode 113 is 8 μm, the gap widths G are 15 μm and the ground planes115 and 117 are 2-3 mm wide. In addition, the optical waveguide 125 ofFIG. 5 is replaced by optical waveguide arms 125A and 125B respectivelydisposed in the substrate 119 respectively underneath the centerelectrode 113 and the ground plane or grounded electrode 117, as shownin FIG. 7. The optical waveguide arms 125A and 125B are formed byselectively applying strips of Ti metal on the surface of the LiNbO₃substrate 119 and diffusing them into the LiNbO₃ substrate at roomtemperature by techniques well known in the art. This is done before theSiO₂ buffer layer 121 and the electrodes 113, 115 and 117 are depositedon the substrate 119.

The optical waveguide arms 125A and 125B are optically coupled togetherat one end to an input waveguide 125C and at a second end to an outputwaveguide 125D to form a Mach-Zehnder interferometer configuration, asshown in FIG. 8.

It should be noted that the FIGS. 5-8 are shown for a Z-cut LiNbO₃substrate, where the Z-axis is normal to the plane of the LiNbO₃substrate. (See insert 118A in FIG. 3.) For an X- or Y-cut LiNbO₃substrate 119 in FIG. 5, the optical waveguide 125 would be similarlyformed in the substrate 119 (as discussed before), but it would belocated between the center electrode 113 and one of the groundedelectrodes 115 and 117. Similarly, for an X- or Y-cut LiNbO₃ substrate119 in FIG. 7, the optical waveguide arms 125A and 125B would besimilarly formed in the substrate 119 (as discussed before), but thewaveguide arm 125A would be located between the center electrode 113 andthe grounded electrode 115, while the waveguide arm 125B would belocated between the center electrode 113 and the grounded electrode 117.

The operation of the embodiment of FIGS. 7 and 8 is substantially thesame as the operation of the embodiment of FIGS. 5 and 6, with thefollowing exceptions. Light at the exemplary 1.3 μm wavelength appliedto optical waveguides 125A and 125B is preferably polarized light withorientation appropriate to the type of substrate and to the type of cut(such as X-cut, Y-cut or Z-cut). For example, an intensity modulatorusing a z-cut LiNbO₃ substrate 119 provides optimal response to verticalpolarized light. The light from the light source 127 is most preferablycw laser light. It may be focused by the lens 129 into the inputwaveguide 125C or may be transmitted via an optical fiber, preferably apolarization-maintaining fiber. The light from the light source 127 isdivided into the optical waveguide arms 125A and 125B of theinterferometer. At the same time, a modulating microwave drive signal,at an arbitrary amplitude of up to the drive voltage V.sub.π and at afrequency in a preselected range, is applied from the microwave source131 to the coplanar waveguide structure 111 (between the input port 152on the center electrode 113 and each of the grounded electrodes 115 and117) and on the same side of the optical modulator as the exemplary 1.3μm light is transmitted into the optical waveguide 125. The centerelectrode 113 also includes an output port 153 for termination of thecoplanar waveguide structure 111. As discussed further below, the drivevoltage V.sub.π is also known as the half-wave voltage V.sub.π, which isthe change in microwave voltage applied to the input port 152 whichresults in phase reversal, i.e. phase shift by a half-wave, i.e. phaseshift by π radians. The half-wave voltage V.sub.π (f) generallyincreases with frequency of the applied microwave signal and istypically on the order of no more than 4 to 5 volts for DC. Thepreselected frequency range can be as high as from 0 Hz up tosubstantially 40 GHz.

The modulating drive signal modulates the light or optical waves in thewaveguide arms 125A and 125B. The light in the two arms 125A and 125B isphase shifted in opposite directions before it is recombined in theoutput waveguide 125D to produce an output beam which contains intensityor amplitude modulation. It is the modulating microwave drive signalwhich modulates the intensity of the light in the interferometer at thefrequency of the microwave drive signal. More particularly, the opticalintensity or amplitude modulation results from an interaction betweenthe optical wave in the optical waveguides 125A and 125B and themicrowave drive signal in the coplanar waveguide structure 111.

The object of velocity matching in the invention is to implement theoptical phase modulator of FIGS. 5 and 6 or the optical intensitymodulator of FIGS. 7 and 8 so as to cause the velocity of the microwavewave to be the same as or substantially equal to the velocity of theoptical wave. That will result in an improved optical response for theoptical modulator. The velocity of the optical wave is the velocity oflight divided by the optical index, while the velocity of the microwavewave is the velocity of light divided by the microwave index.

Three different optical modulator devices were fabricated on substrates119 that were 0.16-0.25 mm thick and 8 mm wide and with the remaininggeometries specified above for the embodiment of FIGS. 7 and 8. Morespecifically, as indicated in FIG. 9, the thicknesses of the electrodes113, 115 and 117 on each of the substrates 119 (and the associatedmicrowave indices) of the three above-noted optical modulators wereabout 11.5 μm (and about 2.44 n_(m)), about 14.6 μm (and 2.37 n_(m)) andabout 18 μm (and about 2.29 n_(m)). The electrode interaction length L₁(FIG. 8) was 24 mm. Finite element calculations indicated that thesegeometries should result in microwave indices n_(m) between 2.2 and 2.4,as shown in FIG. 9. FIG. 9 will now be discussed.

FIG. 9 illustrates a plot of the CPW microwave index n_(m) againstelectrode thickness, and a comparison of theoretical results withexperimental results. More specifically, FIG. 9 shows the optical indexat 2.15, as represented by the horizontal line 141. FIG. 9 also shows acalculated version of a coplanar waveguide microwave (or effective)index of the CPW coplanar mode, as represented by the sloping line 143,as a function of the thickness of the electrodes 113, 115 and 117 forthe above-specified geometries. FIG. 9 also shows three data points 145,147 and 149 for the above-noted three fabricated devices of differentelectrode thicknesses. Note that the microwave index of each of thesedata points 145, 147 and 149 decreases as the associated electrodethickness increases. FIG. 9 shows how well each of the three devices isvelocity matched. Note that the line 141 (representing an optical indexof 2.15) and the line 143 (representing the CPW microwave index vs.electrode thickness of the electrodes 113, 115 and 117) intersect atabout an electrode (113, 115 and 117) thickness of about 20 μm. An exactvelocity match would be at a microwave index of 2.15, which is theoptical index for a wavelength of 1.3 μm. Thus, to obtain an exactvelocity match, the coplanar waveguide microwave (or effective) index143 has to be made equal to the optical index 141. As shown in FIG. 9,this occurs at the point 144 where the lines 141 and 143 intersect.

The coplanar microwave index of the CPW coplanar mode for a givengeometry of the electrode structure is determined by the thickness ofthe electrodes 113, 115 and 117. Therefore, the electrodes 113, 115 and117 must be thick enough so that the coplanar waveguide microwave indexis equal to the optical index in the uniaxial crystal (which is 2.15 inLiNbO₃). However, it should be realized that an exact velocity matchcould be obtained for a different optical index (for a differentwavelength and/or for a different crystal material) by following asimilar procedure.

The optical modulator of the invention can be implemented to operatewith any optical wavelength, but the above-discussed optical modulatorsof the embodiment of FIGS. 5 and 6 and the embodiment of FIGS. 7 and 8were implemented to operate at an optical wavelength of 1.3 μm.

Referring now to FIGS. 10A and 10B, FIG. 10A illustrates the electricaltransmission through the coplanar microwave waveguide structure 111(FIGS. 7 and 8 or FIGS. 5 and 6); and FIG. 10B illustrates thenormalized optical response of the device of FIGS. 5 and 6 or of FIGS. 7and 8 in dB of electrical power as a function of frequency.

FIG. 10A shows that the magnitude of S₂₁ in dB plotted against frequencyover the frequency range from 0 Hz to 40 GHz, where S₂₁ is atransmission parameter from a network analyzer used in the measurement.FIG. 10A basically indicates what the transmission of the opticalmodulator of the invention is in dB as a function of frequency.

FIG. 10B shows the normalized response of the optical modulator of theinvention in dB of electrical power, not optical power, as a function offrequency over the frequency range from 0 Hz to 40 GHz. Experimentaldata points are shown about a theoretical normalized optical responseline 151. Note that the normalized optical response decreases with anincrease in frequency. With a velocity matched optical modulator thedrop off with frequency will be minimized. Ideally a flat response overthe frequency range from 0 Hz to 40 GHz is desired. However, FIG. 10Bshows that the optical modulator of the invention that was tested onlydropped about 7.5 dB over the frequency range from 0 Hz to 40 GHz, whichis a very good response.

Therefore, what has been described in preferred embodiments of theinvention is a broadband, electro-optic modulator which, in anembodiment shown in FIGS. 5 and 6 comprises: a substrate havingsubstrate modes, having electro-optic effects, and having a firstoptical waveguide adapted to receive and transmit light therethrough ina first direction and with a first phase velocity; a buffer layerdisposed on the substrate; and a coplanar waveguide electrode structurehaving a coplanar mode and being disposed on the buffer layer forreceiving an electrical signal propagating therethrough in the firstdirection with a second phase velocity to phase modulate the light inthe optical waveguide at a frequency in the range from 0 Hz up tosubstantially 40 GHz. The substrate has a sufficiently small thicknessso that coplanar-substrate coupling substantially does not occur over adesired frequency bandwidth of operation, and the coplanar waveguideelectrode structure has a sufficiently large thickness so that thesecond phase velocity of the electrical signal is substantially equal tothe first phase velocity. In an embodiment shown in FIGS. 7 and 8, anintensity modulator is produced by adding a second optical waveguidewhich, in combination with the first optical waveguide, forms aninterferometer. The interaction of the electrical signal with theoptical signal produces an intensity-modulated optical beam. Thisexemplary modulator is described further in G. K. Gopalakrishnan et al,"Electrical Loss Mechanisms in Travelling Wave LiNbO₃ OpticalModulators," Electronics Letters, Vol.28, No.2, pp. 207-208 (16 Jan.1992) and in G. K. Gopalakrishnan et al, "40 GHz, Low Half-Wave VoltageTi:LiNbO₃ Intensity Modulators," Electronics Letters, Vol.28, No.9, pp.826-827 (23 Apr. 1992), which articles are incorporated herein byreference.

The above-described exemplary intensity modulator is a traveling wavemodulator, as distinguished from a lumped element modulator. Asdiscussed in L. M. Johnson, "Relative Performance of Impedance-MatchedLumped-Element and Traveling-Wave Integrated-Optical Phase Modulators,"(IEEE Photonics Technology Letters, Vol. 1, No. 5, May 1989, pp.102-104), which is incorporated herein by reference, a lumped elementmodulator is modeled as and characterized by resistance and capacitanceand by an impedance which varies with electrical frequency. A travelingwave modulator is modeled as and characterized by an impedance which isrelatively independent of electrical frequency. Lumped elementmodulators are typically bound with wires to other components; travelingwave modulators are typically part of microwave transmission lines. Theformer type of modulator is typically more effective at lowerfrequencies (say, below 1 megahertz (MHz)) than at higher frequencies,and the latter type of modulator is typically more effective at higherthan at lower frequencies.

The above-described exemplary intensity modulator is consideredbroadband, since it has good response over a preselected frequency rangeon the order of tens of GHz.

Having described the invention in general, the following example isgiven as a particular embodiment thereof and to demonstrate the practiceand advantages thereof. It is understood the example is given by way ofillustration and is not intended to limit the specification or theclaims to follow in any manner. This example is discussed further in G.K. Gopalakrishnan et al, "A LiNbO₃ Microwave-Optoelectronic Mixer withLinear Performance," 1993 IEEE MTT-S Digest, Conference Proceedings,International Microwave Symposium, Atlanta, Ga., pp. 1055-1058; G. K.Gopalakrishnan et al, "Microwave-Optical Mixing in LiNbO₃ Modulators,"IEEE Transactions on Microwave Theory and Techniques, Vol. 41, No.12,pp. 2383-2391 (December 1993), which articles are incorporated herein byreference.

Referring now to FIG. 11, the above described exemplary modulators M₁and M₂ may include biasing means 180 and 190, respectively. For each ofthe modulators M₁ and M₂, a DC bias voltage S_(dc1) and S_(dc2) isapplied to a bias T-connector 200 and 210, respectively. An isolator 220and 230 is coupled between the microwave signal producer 20 and the biasT-connector 200 and 210, respectively. As a result, the biasing means180 and 190 produce biased microwave signals S₁ ' and S₂ ' for input tothe input ports 152₁ and 152₂ of modulators M₁ and M₂, respectively.

Referring back to FIGS. 7 and 8, the modulators M₁ and M₂ are designedfor operation at an optical wavelength of 1.3 μm with the characteristicimpedance of the CPW line (Z_(D)) being approximately 35 ohms (Ω). Theyare fabricated as follows: The electrodes 113, 115 and 117 arepreferably made of gold and have thicknesses of 15 and 18 μm,respectively. The electrode 113 has a width of 8 μm. The gap width Gbetween the center electrode 113 and each of the grounded electrodes 115and 117 is selected to be 15 μm, while the grounded electrodes 115 and117 are selected to be about 2-3 mm wide. The substrate 119 has a widthof 8 mm and a thickness of from 0.2 to 0.15 mm for modulators M₁ and M₂,respectively. The electrode interaction length L₁ was 24 mm. Thesubstrate 119 is coated with a 0.9 μm SiO₂ buffer layer 121.

As so fabricated, the modulators M₁ and M₂ have DC half-wave voltagesV.sub.π of 4.2 and 5 Volts, respectively. When biased for maximumtransmission, the optical insertion loss through each modulator isapproximately 4 dB.

The above-described exemplary intensity modulator is considered a lowdrive voltage modulator since the DC drive voltage V.sub.π (DC) is belowabout 5 Volts. The significance of the modulator being a low drivevoltage modulator is discussed further below.

From a fit to the electrical response of the modulator comprisingconductor, dielectric and radiative losses and considering themicrowave-optical index mismatch (0.06 for modulator M₁ and 0.128 formodulator M₂) obtained from a fit to the optical response, the modeledoptical responses of the modulators M₁ and M₂ are shown in FIG. 12.Referring back to FIG. 11, the model also takes into considerationreflections that occur at the transition from the 35Ω CPW line to the50Ω load 240 and 250 at the output ports 153₁ and 153₂, respectively.Here, a concern is to characterize traveling wave modulators in terms oftheir mixing efficiency. A fundamental mixer parameter that reflectsmixing efficiency is its RF conversion loss. As will be discussedfurther later, the RF half-wave voltage V.sub.π (f) of the modulators M₁and M₂, adequately characterizes the mixing efficiency of the modulatorsM₁ and M₂. The RF half-wave voltage V.sub.π (f) of the modulators M₁ andM₂ may be obtained from the optical response of the modulators M₁ and M₂by using the following expression: ##EQU3## where V.sub.π (DC) is thehalf-wave voltage of the modulator at DC, Z₀ =50Ω is the characteristicimpedance of the line, Z_(D) is the characteristic impedance of the CPWelectrode of the device, and OR is the optical response of the modulatorin electrical decibels. Also shown in FIG. 12 are the modeled RFhalf-wave voltages V.sub.π (f) of M₁ and M₂ as a function of frequencyf.

Referring back to FIG. 11, the above-described exemplary pair ofinterferometric modulators M₁ and M₂ are cascaded in series, anddown-conversion is demonstrated. This is done by applying a single-toneRF input S₁ of angular frequency ω₁ to one modulator M₁, the modulatedoptical output L_(m1) of which is mixed with a larger LO localoscillator pump signal S₂ of angular frequency ω₂ applied to the othermodulator M₂. The modulator M₂ produces a second modulated opticalsignal L_(m2) having an angular frequency component at IF, i.e. thedifference |ω₁ -ω₂ | between the angular frequencies ω₁ and ω₂. Thesecond modulated optical signal L_(m2) is detected with a photoreceiver260, the output of which is applied to a spectrum analyzer 270 having a50Ω load. The second modulated optical signal L_(m2) produced by themodulator M₂ also has an angular frequency component at the sum ω₁ +ω₂of the angular frequencies ω₁ and ω₂, but his component is not asrelevant to down-conversion applications as the IF component.

The actual signals that take part in the mixing are optical signals (notshown) at the RF (ω₁) and LO (ω₂) frequencies, derived from electricalinputs S₁ ' and S₂ ', respectively, applied to modulators M₁ and M₂,respectively. To efficiently down-convert these to a desired IF, opticalsignals at both the RF and LO frequencies must be highly modulated, and,hence, to prevent leakage of power to other unwanted frequencies, bothmodulators M₁ and M₂ should be biased at quadrature.

In this configuration, the intended application of the cascadedmodulator pair M₁ --M₂ is for antenna remoting. Antenna remoting usuallyrefers to an arrangement of source, fiber coupled remote (from thesource and often from electrical power) modulator, and fiber coupleddetector (usually located with the source). The RF signal from thedetector is then usually electrically down-converted in a mixer andprocesses.

Upon application of this invention to antenna remoting, the RF signal isfed to one modulator, say, M₁ and the local oscillator LO to the other,say, M₂, with the modulator receiving the RF signal considered theremote modulator. The mixing process provides down-conversion, and onlythe IF needs to be detected. For such an application, one modulator M₁can be used to transmit the relatively low-power RF-modulated opticalsignal L_(m1) from a remote location over an optical fiber. The RFsignal can then be down-converted by mixing the RF-modulated opticalsignal L_(m1) with a relatively large LO pump signal S₂ applied to theother modulator M₂. Electrical mixing is no longer needed and the weakpoints of conventional microwave mixers, described earlier, areeliminated. The advantages of this approach for antenna remoting arethat the IF is obtained directly at the detector, thus reducing demandson the detector, since the detector does not need to be broadband, andthat the mixing is accomplished optically with broad bandwidth, on theorder of tens of GHz, and with infinite port-to-port isolation.

This tandem arrangement of modulators M₁ and M₂ also allows formultiplication of RF signals applied to each modulator, and toup-conversion of RF signals applied to each modulator.

For a single Mach-Zehnder interferometer modulated by electrical RFsignal V·sin(ωt) and having half-wave voltage V.sub.π, the normalizedratio of output optical power P_(o) to input optical power P_(i) is##EQU4## where φ₀ is the intrinsic phase bias. In this derivation, eachinterferometer M₁ and M₂ is biased at quadrature, so φ₀ =90°.

If RF input signals S₁ =V₁ ·sin(ω₁ t) and S₂ =V₂ ·sin(ω₂ t) are appliedto modulators M₁ and M₂, respectively, each biased at quadrature, thenusing equation (5), the ratio of the output optical power P_(o) of thesignal L_(m2) to the input optical power P_(i) of the source light L_(i)is ##EQU5## where T_(D) is the coupling and optical transmission lossesof the system and V.sub.π1 (ω₁) and V.sub.π2 (ω₂) are the RF half-wavevoltages of modulators M₁ and M₂ at angular frequencies ω₁ and ω₂,respectively. Rewriting and expanding equation (6) and neglecting termshigher than third order, we get: ##EQU6## where J_(n) is the Besselfunction of order n, X₁ =(π·V₁)/(V.sub.π1 (ω₁)), and X₂=(π·V₂)/(V.sub.π2 (ω₂)) .

From Equation (7), for X₁ <<1 and X₂ <<1, neglecting power terms higherthan third order, we get ##EQU7## In equation (8), the terms associatedwith the third harmonic frequencies (3ω₁ and 3ω₂) are small compared tothe fundamental, sum and difference terms and hence can be neglected. Itcan be seen from the above derivation of equation (8) that the thirdorder IM frequency terms (2ω₁ -ω₂ and 2ω₂ -ω₁) are not present when eachinterferometer of the cascaded pair M₁ --M₂ is biased at quadrature. Inconventional mixers, if ω₁ ≈ω₂, then the IM frequencies would be veryclose to the signal frequencies and thus could not be filtered out.However, as shown, the present invention has low IM distortion.Furthermore, the power associated with the IM frequencies isproportional to the cube of the input voltages V₁ and V₂, so the powerincreases rapidly with input power thereby limiting the dynamic range ofthe mixer 10. The cascaded Mach-Zehnder interferometric pair M₁ --M₂,when biased at quadrature, does not suffer from such drawbacks and henceis an attractive candidate for microwave-optoelectronic mixing.

The electrical power delivered by the photodetector 260 to the 50Ω loadfor angular frequencies ω₁ -ω₂ and ω₁ +ω₂ can be evaluated using thefollowing expression: ##EQU8## where L(ω) collectively representsangular frequency dependent losses such as detector roll-off and cablelosses that limit the performance of the fiber-optic link, and I_(DC) isthe detected DC photocurrent.

To demonstrate the effect of quadrature biasing on IM distortion, themodulators M₁ and M₂ were biased off-quadrature, and equal amplitudeinput signals at f₁ =15.52 GHz (RF) and f₂ =15.56 GHz (LO) were appliedto the modulators M₁ and M₂, respectively. The output spectrum, inaddition to the signal terms f₁ and f₂, contains the following beatsignals: the desired IF difference signal at f₂ -f₁ =40 MHz, and theundesired intermodulation signals at 2f₁ -f₂ =15.48 GHz and 2f₂ -f₁=15.6 GHz. This is illustrated in FIG. 13, where the spectra of thesignal and IM frequencies are shown. However, when each device M₁ and M₂of the cascaded pair is biased at quadrature, the IM signals disappear,resulting in optical signals at just the signal and differencefrequencies, as illustrated in FIG. 14. Since the mixer 10 conservesoptical power, the less the power of output IM signals, the greater thepower of output sum and IF signals, and therefore the more efficient themixer 10 is. The mixer 10 in which modulators M₁ and M₂ are biased atquadrature is a more efficient mixer. In comparing FIGS. 13 and 14, itcan be seen that more power is output in the sum and IF frequencies whenthe modulators M₁ and M₂ are biased at quadrature.

To demonstrate that the IF can be anywhere in the broadband DC to 40 GHzfrequency range, the experiment was extended to higher frequencies.Signals S₁ and S₂ at frequencies f₁ =15 GHz and f₂ =6 GHz were appliedto the modulators M₁ and M₂, respectively and the mixer 10 produced theIF difference signal at 9 GHz.

For input signals at f₁ =15.52 GHz (RF) and f₂ =15.56 GHz (LO), FIG. 15shows the variation of detected IF power at 40 MHz with input RF powerfor a detected DC photocurrent of 0.05 mA; the LO pump power was variedfrom -20 to +20 dBm (0.01 mW-100 mW). At these values of RF and LO,V.sub.π1 (f₁)=7.7 V and V.sub.π2 (f₂)=8.9 V. Since the IF was at 40 MHz,L(ω) is negligible. Here, with the LO pump power at +20 dBm, theconversion loss is approximately 66 dB. At 40 GHz (f₁ =40.02 GHz, f₂=40.06 GHz; V.sub.π1 (f₁)=10.4 V and V.sub.π2 (f₂)=14.2 V), thecorresponding computed results are shown in FIG. 16 and the conversionloss in this case for a pump power of +20 dBm is about 72 dB. Here, therelatively large conversion loss is attributable in part to the opticalinsertion loss of the interferometric pair M₁ --M₂. We were limited toapproximately 12 mW of optical power from the laser at the input to thefirst modulator M₁. At the output, for a detected DC photocurrent of0.05 mA, the power striking the detector 260 was 0.1 mW (assuming adetector responsivity of 0.5 Amp/Watt) indicating an optical link lossin excess of 20 dB. Here, the modulators M₁ and M₂ biased at quadraturecontribute to 14 dB (approximately 7 dB each) of this loss, and couplinglosses account for the rest.

The conversion loss of the cascade interferometric pair M₁ --M₂ would besignificantly lower if the power on the photodetector 260 were larger.This may be accomplished by employing more powerful lasers 32, or byemploying erbium doped fiber amplifiers (EDFA). In this context, we noteagain that the optical LO pump signal in our experiments was derived byapplying a large electrical input to the modulator. If this input wereincreased further, then beyond a certain point, compression of theoutput would occur due to sinusoidal modulator response. In other words,for amplitude greater than the half-wave voltage V.sub.π (f), themodulation would swing over more than a half-wave. However, with erbiumdoped fiber amplifiers (EDFA) currently becoming available at an opticalwavelength of 1.5 μm, a better way to generate this optical pump signalwould be, as shown in FIG. 19, to employ an EDFA 280 to amplify themodulated optical LO pump signal L_(m1) which could then be mixed with alower RF signal S₂ to accomplish down-conversion. Alternatively, asshown in FIG. 20, an EDFA 280 could be employed to amplify the output IFsignal.

The mixing efficiency of the interferometric pair for RF down-conversionis dependent on the strengths of the modulated optical RF and LOsignals. It is, therefore, of interest to study optical modulation atthe RF and LO signal frequencies. Towards this end, we applied twoindependent equal amplitude signals S₁ and S₂ spaced 40 MHz apart to thecascaded interferometric modulator pair M₁ --M₂ biased at quadrature;the frequencies were centered around 15.5 GHz. For the modulating RFinput corresponding to modulator M₁, we show in FIG. 17 the variation ofthe detected signal power as a function of input RF power, for aphotodetector 260 current of 0.05 mA. for this measurement, L(ω)=-3 dB.As shown, this data is in fairly good agreement with theory with theoutput increasing linearly with input for power levels that do not causecompression. Also shown, the computed response at 40 GHz assuming thesame DC photocurrent and the same L(ω). In actuality, however, thesignal strength at 40 GHz would be smaller than that predicted becauseL(ω) would be worse than -3 dB.

The significance of the exemplary modulators M₁ and M₂ having low drivevoltage V.sub.π is in relation to conversion loss. The conversion lossis a fundamental mixer parameter employed to define the mixingefficiency. Conversion loss (CL) is defined as follows:

RF CL (dB)=detected IF power (dBm)-input RF power (dBm). (10)

The lower the conversion loss, the larger the detected down-converted IFsignal strength, and hence the better the signal to noise ratio. For thefirst signal S₁ being the RF signal, and the second signal S₂ being theLO signal, equation (9) becomes: ##EQU9## In equation (11), the input RFpower is contained in the J₁ (X_(RF)) term, and the LO pump power iscontained in the J₁ (X_(LO)) term. The laser power is contained in theI_(DC) term. Employing the small-signal argument for the Besselfunction, it can be seen that the detected IF (|RF-LO|) output power isinversely proportional to the fourth power of the frequency dependenthalf-wave voltage V.sub.π of the modulators M₁ and M₂, and is directlyproportional to the square of the detected DC current. Thus, theconversion loss of the mixer 10 can be decreased by either increasingthe detected optical signal power, such as by employing high powerlasers or optical amplifiers, or by employing modulators with lowerhalf-wave voltages. A small decrease in half-wave voltage V.sub.π 33%results in the detected power being multiplied by a disproportionallylarge factor. For example, a decrease in half-wave voltage V.sub.π by33% results in the detected power being multiplied by a factor of (3/2)⁴=2.5. The modulator discussed in D. W. Dolfi et ano., "40 GHzElectro-Optic Modulator with 7.5 V Drive Voltage," Electronics Letters,Vol 24, No. 9, pp. 528-529. (April, 1988) discusses a modulator havingDC half-wave voltage of 7.5 V. As discussed above, the exemplarymodulators M₁ and M₂ herein have DC half-wave voltage of about 5 V.Therefore, the mixer disclosed herein has detected power of about 2.5times the detected power of the mixer disclosed in Johnson '086 usingthe modulators of Dolfi, supra.

Referring now to FIG. 18, the conversion loss in decibels is plottedagainst the RF Half-Wave voltage V.sub.π (f)(RF). As shown, theconversion loss increases rapidly as the RF half-wave voltage increases.The modulator discussed in Dolfi (supra) exhibits an RF half-wavevoltage of 24 V. Thus, for a detected DC photocurrent of 1 milliAmp (mA)and a LO pump power of 20 dBm (100 milliWatts (mW), a cascade-connectedpair of such modulators would provide a conversion loss of about 55 dB.The above-described exemplary modulator has an RF half-wave voltage of10.4 V, and so the mixer shown in FIGS. 1 and 11 would have a conversionloss of about 43 dB, 12 dB better than Dolfi. A further improvement of20 dB could be obtained if the detected photo-current could be increasedto 10 mA with a more powerful laser or optical amplifier, as discussedfurther below. Overall, a conversion loss of less than about 40 dBprovides a practical mixer. In systems with higher conversion losses,the modulated signal would be masked by noise.

For a detected photo-current of 0.05 mA, the measured signal-to-noiseratio of the mixer described herein in which amplifiers are not employedat the output is about 37 dB for IF at 9 GHz, and about 40 dB for IF at40 MHz. These measurements reflect the noise of the spectrum analyzer270.

In a fiber optic link, baseband information is transmitted over anoptical fiber from the transmitter end to the receiver end, whichrecovers the baseband information.

Referring now to FIG. 21, an exemplary fiber optic link transmissionsystem 290 is based on the above-described mixer (FIG. 1). Thisexemplary transmission system 290 is discussed further in G. K.Gopalakrishnan et al, "A Fiber-Optic Link for Microwave SubcarrierTransmission and Reception," 1995 IEEE MTT-S Digest, ConferenceProceedings, International Microwave Symposium, Orlando, Fla.,pp.1165-1168 (May 15-19, 1995).

The fiber link transmission system 290 includes first and second opticalinterferometric modulators M₁ and M₂, each preferably being broadband,traveling-wave intensity modulators having DC drive voltage below about5 V, and most preferably Ti:LiNbO₃ traveling-wave intensity modulatorsas described above. The first modulator M₁ is driven by polarized lightL_(i), and most preferably polarized cw laser light L_(i) produced bydevice 30.

A baseband source 300 produces the baseband information signal S_(i). Anexemplary baseband signal S_(i) used for test purposes is a 50 Mbit/snon-return-to-zero (2²³ -1) pseudo-random bit stream from the datatransmitter of a bit error rate (BER) test set which is amplitude shiftkeyed (ASK). The baseband information signal S_(i) is preferably, butnot necessarily digital, and is not necessarily modulated. Other formsof modulation and especially pulse code modulation known in the art maybe used, such as frequency shift keyed (FSK), phase shift keyed (PSK),quaternary phase shift keyed (QPSK), and quadrature amplitude modulation(QAM). The baseband information signal S_(i) is applied to the IF portof an electrical microwave mixer M_(o) ^(e) and upconverted to themicrowave frequency band by application of a microwave signal S_(rf) offrequency f_(rf) produced by an RF synthesizer 310 and applied to the RFport of the mixer M_(o) ^(e). The frequency f_(rf) for the microwavesignal S_(rf) is the subcarrier frequency and exemplary subcarrierfrequencies f_(rf) are 9 GHz and 16 GHz. Microwave modulators M_(o) ^(e)with performance to 75 GHz have been demonstrated, so the techniqueadvanced herein could easily be extended to higher subcarrierfrequencies.

The electrical output of the mixer M₀ ^(e) is applied to the firstoptical modulator M₁ as signal S₁ ". Although the mixer M₀ ^(e) outputcould be applied to the first optical modulator M₁ by standard microwavetransmission means, it could also be transmitted in free space to themodulator M₁. Thus, the mixer M₀ ^(e) and modulator M₁ could be remotefrom each other and the transmission system 290 could be used inwireless communications.

The first optical modulator M₁ responsive to the signal S₁ " and thesource light L_(i) responsively produces a first modulated opticalsignal L_(m1). This first modulated optical signal L_(m1) is responsiveto the upconverted signal S₁ " and so the process described so farinvolves subcarrier modulation with subcarrier frequency f_(rf)determined by the microwave signal S_(rf).

The upconverted signal L_(m1) is transmitted, preferably but notnecessarily by polarization-maintaining means 40, and most preferablypolarization-maintaining optical fiber (PMF) 40, to the second modulatorM₂. The second modulator M₂ is disposed apart from and may be somedistance from the first modulator M₁. The second modulator M₂ is cascadecoupled to the first modulator M₁ and responsive to the first modulatedsignal L_(m1) and to a second electrical microwave signal S₂ " forproducing a second modulated optical signal L_(m2) having a componentwith frequency at the difference between the frequencies of theelectrical signals S₁ " and S₂ " respectively. The second electricalmicrowave signal S₂ " is an LO pump signal produced by an LO synthesizer320. The second modulator M₂ thus performs downconversion. Inparticular, the subcarrier frequency f_(IF) of the second modulatedoptical signal L_(m2) is the difference between f_(rf) and the frequencyof S₂ ", say, f_(IF) =200 MHz.

The second modulated optical signal L_(m2) is detected by aphotoreceiver 260 with exemplary bandwidth of 300 MHz. Photoreceivers260 with bandwidths as high as 10 GHz are presently available and couldjust as readily be used. The electrical output of photoreceiver 260 isthen processed to recover the baseband data as a signal S_(o).

The photoreceiver 260 need only have sufficient bandwidth to detect theIF subcarrier and recover the baseband information. Because ofdownconversion by the second optical modulator M₂, it is not necessaryfor the photoreceiver 260 to be of high enough speed to detect thesubcarrier frequency f_(rf). Therefore, the problems associated withhigh speed photodetectors are avoided.

An exemplary system for recovering the baseband data includes a 150 MHzhigh-pass filter (HPF) 330 and an amplifier 340 for amplifying thesignal by about 30 dB. The output of the amplifier 340 and an IF signalat the frequency f_(IF) produced by an IF synthesizer 350 are mixed byan electrical mixer M₃ ^(e) to produce the output information signalS_(o).

Optionally, to avoid potential subcarrier and clock recovery, the RFsynthesizer 310, LO synthesizer 320, and IF synthesizer 350 are phaselocked to a 10 MHz reference provided by a reference source 355.

For purposes of testing the apparatus 290, the output signal S_(o) wasfiltered with a 50 MHz low pass filter 360 and applied to an errordetector 370 of the BER test set.

Referring now to FIG. 22, The BER as a function of received opticalpower is shown for two different carrier frequencies (9 and 16 GHz).Error free transmission (corresponding to a BER of less than 10⁻⁹ wasachieved at received optical powers of ≧-30.5 dBm at 9 GHz and ≧-28.5dBm at 16 GHz. When the carrier frequency is increased from 9 to 16 GHz,there is a power penalty of about 2 dB in achieving error freetransmission. This can be attributed to roll-off in the frequencyresponse of the modulators M₁ and M₂.

The subcarrier frequency f_(rf) can be readily scaled up with minimalpower loss. Such higher frequency has benefits in terms of ability tocarry more data.

The above-described embodiment of a fiber optic link transmission systemcould be readily extended to a systems utilizing multiple subcarrierchannels. Referring now to FIG. 23, each baseband information signalS_(i1), S_(i2), . . . is applied to the IF port of a corresponding mixerM₀,1^(e), M₀,2^(e), . . . and upconverted to the microwave frequencyband by application of a corresponding microwave RF signal S_(rf1),S_(rf2), . . . to the RF port of the corresponding mixer to produce amicrowave modulated signal, i.e. baseband information signal S_(i1),S_(i2), . . . modulated by microwave subcarrier signal S_(rf1), S_(rf2),. . . The microwave modulated output signals of the mixers M₀,1^(e),M₀,2^(e), . . . are applied by standard microwave transmission means, bywireless transmission in free space, or other means known in the art, tocorresponding first optical modulators M₁,1, M₁,2, . . . ascorresponding modulated signals S₁,1 ", S₁,2 ", . . . The first opticalmodulators M₁,1, M₁,2, . . . , responsive to corresponding modulatedsignals S₁,1 ", S₁,2 ", . . . , produce corresponding first modulatedoptical signals L_(m1),1, L_(m1),2, . . . which are applied, preferablyby polarization-maintaining optical fiber, to an optical combiner 380.The optical combiner 380, preferably having low cross-interference,responsive to the first modulated optical signals L_(m1),1, L_(m1),2, .. . , produces a combined first modulated optical signal L_(m1) '. Thiscombined first modulated optical signal L_(m1) ' includes the basebandinformation of baseband information signals S_(i1), S_(i2), . . .multiplexed on separate subcarrier subchannels determined by thesubcarrier signals S_(rf1), S_(rf2), . . .

The combined first modulated optical signal L_(m1) ' is transmitted,preferably by polarization-maintaining optical fiber, to a splitter 390,which produces received first modulated optical signals L_(m1),1 ",L_(m1),2 ", . . . , and transmits them to corresponding second opticalmodulators M₂,1, M₂,2, . . . Each second optical modulator M₂,1, M₂,2, .. . selects a subchannel by application of a corresponding LO microwavesignal S₂,1 ", S₂,2 ", . . . to produce a corresponding second opticalmodulated signal L_(m2),1, L_(m2),2, . . . Some or all of the secondoptical modulated signals L_(m2),1, L_(m2),2, . . . are detected by acorresponding photoreceiver 260₁, 260₂, . . . and processed to recoverthe data on the selected subchannel.

Alternatively, the receiver end of the system 290 (FIG. 21), rather thanthe receiver end 400, could be utilized, with the microwave LO signal S₂" tuned to select the desired channel.

Referring now to FIGS. 21 and 23, in an alternative configuration forthe input end of a fiber optic link transmission system utilizingmultiple subchannels, microwave modulated output signals of the mixersM_(o),1^(e), M_(o),2^(e), . . . are combined by a microwave combiner(not shown) of sufficient bandwidth, the output of which is applied assignal S₁ " to the first optical modulator M₁. The first modulatedoptical signal L_(m1) now includes the baseband information of basebandinformation signals S_(i1), S_(i2), . . . multiplexed on separatesubcarrier subchannels determined by the subcarrier signals S_(rf1),S_(rf2), . . . , just as the combined first modulated optical signalL_(m1) ' does. The first modulated optical signal L_(m1) can beprocessed as the combined first modulated optical signal L_(m1) 'described above is to recover the data on a selected subchannel.

Referring now to FIG. 24, in an optical fiber link transmission system410, an optical modulator M₀ can be used to modulate the basebandinformation signal S_(i). The optical modulator M₀ is preferably of thesame type as optical interferometric modulators M₁ and M₂ describedabove.

The baseband information signal S_(i) is applied to the opticalmodulator M₀ which, responsive to source light L_(i) and the basebandsignal S_(i) produces a modulated optical signal L_(m0). This signalL_(m0) includes the source light L_(i) modulated by the baseband signalS_(i). The modulated optical signal L_(m0) is applied to the firstoptical modulator M₁. The first optical modulator M₁, responsive to themodulated optical signal L_(m0) and the subcarrier signal S_(rf)produces the first modulated optical signal L_(m1). The first modulatedoptical signal L_(m1) is a subcarrier multiplexed signal and is furtherprocessed as in device 290 (FIG. 21).

The fiber optic transmission system 410 utilizes a broadband modulatorM₀ having a much wider bandwidth than an electrical mixer M_(o) ^(e) ofsystem 290. Therefore, this system 410 is capable of processing basebandinformation signals S_(i) with high information content and high bitrate.

Referring now to FIG. 25, a fiber optic link transmission systemutilizing multiple subcarrier channels has an optical interferometricmodulator M₀,1, M₀,2, . . . for each channel. Each baseband informationsignal S_(i1), S_(i2), . . . is applied to the corresponding opticalmodulator M₀,1, M₀,2, . . . which, responsive to it and to source lightL_(i), produces a corresponding modulated optical signal L_(m0),1,L_(m0),2, . . . Each modulated optical signal L_(m0),1, L_(m0),2, . . .includes the source light L_(i) modulated by the corresponding basebandsignal S_(i1), S_(i2), . . . Each modulated optical signal L_(m0),1,L_(m0),2, . . . is applied to a corresponding first opticalinterferometric modulator M₁,1, M₁,2, . . . Each first opticalinterferometric modulator M₁,1, M₁,2, . . . responsive to thecorresponding modulated optical signal L_(m0),1, L_(m0),2, . . . and toa corresponding subcarrier signal S_(rf1), S_(rf2), . . . produces afirst modulated optical signal L_(m),1,1, L_(m1),2, . . . Each firstmodulated optical signal L_(m1),1, L_(m1),2, . . . is a subcarriermultiplexed signal and is further processed as described above and shownin FIG. 23.

The foregoing descriptions of the preferred embodiments are intended tobe illustrative and not limiting. Numerous modifications and variationscan be made without departing from the spirit or scope of the presentinvention as defined in the appended claims. Within the scope of theappended claims, the invention may be practiced otherwise than asspecifically described.

What is claimed is:
 1. A transmission system responsive to an electricalinput information signal comprising:(a) means for modulating the inputsignal with a microwave subcarrier signal to produce a subcarriermodulated signal; (b) a first interferometric modulator responsive to asource light and to the subcarrier modulated signal for producing afirst modulated optical signal; (c) a second interferometric modulatorcascade coupled to said first modulator and being responsive to thefirst modulated optical signal and to a microwave local oscillatorsignal for producing a second modulated optical signal; (d) meanscoupled between said first and second modulators for transmitting thefirst modulated optical signal from said first modulator to said secondmodulator; and (e) means responsive to the second modulated opticalsignal for producing an output signal.
 2. The system of claim 1, saidmodulation means (a) comprising means for modulating a pulse codemodulated digital signal.
 3. The system of claim 1, further comprisingmeans for wireless transmission of the subcarrier modulated signalproduced by said modulation means (a) to said first interferometricmodulator (b).
 4. The system of claim 1, each of said first and secondmodulators (b) and (c), respectively, comprising a broadband, travelingwave intensity modulator having DC drive voltage below about 5 Volts. 5.The system of claim 1 further comprising means for biasing each of saidfirst and second modulators (b) and (c), respectively, at quadrature. 6.The system of claim 1, said second modulator (c) comprising a modulatordisposed at a predetermined distance from said first modulator (b). 7.The system of claim 1, said transmitting means (d) comprisingpolarization-maintaining means.
 8. The system of claim 7, saidtransmitting means (d) further comprising a polarization-maintainingoptical fiber.
 9. The system of claim 1, said output signal producingmeans (e) comprising:a photoreceiver responsive to the second modulatedoptical signal; and a mixer responsive to the output of saidphotoreceiver and to a microwave IF signal for producing the outputsignal.
 10. A transmission system responsive to a plurality ofelectrical information input signals comprising:(a) a plurality ofmeans, each corresponding to an input signal, for modulating thecorresponding input signal with a corresponding microwave subcarriersignal to produce a corresponding subcarrier modulated signal; (b) aplurality of first interferometric modulators, each corresponding to asubcarrier modulated signal, each responsive to a source light and tothe corresponding subcarrier modulated signal for producing acorresponding first modulated optical signal; (c) an optical combinerresponsive to the plurality of first modulated optical signals producedby said plurality of first interferometric modulators for producing acombined first modulated optical signal, the combined first modulatedoptical signal having a plurality of subchannels corresponding to saidplurality of subcarrier modulated signals; (d) a second interferometricmodulator cascade coupled to said combiner and being responsive to thecombined first modulated optical signal and to a microwave localoscillator signal for producing a second modulated optical signal; (e)means coupled between said combiner and said second optical modulatorfor transmitting the combined first modulated optical signal from saidcombiner to said second modulator; and (f) means responsive to thesecond modulated optical signal for producing an output signal.
 11. Thesystem of claim 10 further combining means for selectively tuning themicrowave local oscillator signal so as to select at least one of theplurality of subcarrier channels.
 12. A transmission system responsiveto an electrical input information signal comprising:(a) aninterferometric modulator responsive to a source light and to the inputsignal for producing a modulated optical signal; (b) a firstinterferometric modulator cascade coupled to said interferometricmodulator and being responsive to the modulated optical signal and to amicrowave subcarrier signal for producing a first modulated opticalsignal; (c) a second interferometric modulator cascade coupled to saidfirst modulator and being responsive to the first modulated opticalsignal and to a microwave local oscillator signal for producing a secondmodulated optical signal; (d) means coupled between said interferometricmodulator and said first modulator for transmitting the modulatedoptical signal from said interferometric modulator to said firstmodulator; (e) means coupled between said first and second modulatorsfor transmitting the first modulated optical signal from said firstmodulator to said second modulator; and (f) means responsive to thesecond modulated optical signal for producing an output signal.
 13. Thesystem of claim 12, said interferometric modulator (a) comprising amodulator for modulating a pulse code modulated digital signal.
 14. Thesystem of claim 12, each of said interferometric modulators (a), (b) and(c) comprising a broadband, traveling wave intensity modulator having DCdrive voltage below about 5 Volts.
 15. The system of claim 12 furthercomprising means for biasing each of said interferometric modulators(a), (b) and (c) at quadrature.
 16. The system of claim 12, said secondmodulator (c) comprising a modulator disposed at a predetermineddistance from said first modulator (b).
 17. The system of claim 12, saidtransmitting means (d) and (e) each comprising polarization-maintainingmeans.
 18. The system of claim 17, said transmitting means (d) and (e)each further comprising a polarization-maintaining optical fiber. 19.The system of claim 12, said output signal producing means (e)comprising:a photoreceiver responsive to the second modulated opticalsignal; and a mixer responsive to the output of said photoreceiver andto a microwave IF signal for producing the output signal.